Transmitter power detection method

ABSTRACT

A method of detecting a power level of an RF signal includes driving an internal node of a power detection circuit to a DC node voltage level, generating, based on the DC node voltage level, a first component of an output voltage on an output node of the power detection circuit, receiving the RF signal on an input node of the power detection circuit, dividing the RF signal to generate a modulation signal on the internal node, and generating, by at least partially rectifying the modulation signal, a second component of the output voltage on the power detection circuit output node.

PRIORITY CLAIM

The present application is a divisional of U.S. application Ser. No.16/133,207, filed Sep. 17, 2018, which is a divisional of U.S.application Ser. No. 15/682,918, filed Aug. 22, 2017, now U.S. Pat. No.10,079,583, issued Sep. 18, 2018, which claims the priority of U.S.Provisional Application No. 62/427,592, filed Nov. 29, 2016, which areincorporated herein by reference in their entireties.

BACKGROUND

Radio frequency (RF) transmitters often are designed to meet industrystandards that facilitate desired communications and avoid undesiredinterference by maintaining compatibility with other devices. To ensuresuch compatibility, industry standards typically include provisionsdirected to transmitter power levels.

In some RF transmitter circuit applications, an output stage includes apower amplifier that is part of an integrated circuit (IC) chip.Transmitter components can include additional IC circuits as well ascomponents such as antennae located externally to the IC chip.

BRIEF DESCRIPTION OF THE DRAWINGS

Aspects of the present disclosure are best understood from the followingdetailed description when read with the accompanying figures. It isnoted that, in accordance with the standard practice in the industry,various features are not drawn to scale. In fact, the dimensions of thevarious features may be arbitrarily increased or reduced for clarity ofdiscussion.

FIG. 1 is a diagram of a transmitter circuit, in accordance with someembodiments.

FIG. 2 is a diagram of a power detection circuit, in accordance withsome embodiments.

FIG. 3 is a diagram of a reference voltage circuit, in accordance withsome embodiments.

FIG. 4 is a diagram of a comparator circuit, in accordance with someembodiments.

FIG. 5 is a timing diagram of voltage and current signals at variousnodes of the power detection circuit of FIG. 2, in accordance with someembodiments.

FIG. 6 is a diagram of a transfer function of the power detectioncircuit of FIG. 2, in accordance with some embodiments.

FIGS. 7A and 7B are timing diagrams of voltage signals at various nodesof the transmitter circuit of FIG. 1, in accordance with someembodiments.

FIG. 8 is a flowchart of a method of detecting a power level of an RFsignal, in accordance with some embodiments.

DETAILED DESCRIPTION

The following disclosure provides many different embodiments, orexamples, for implementing different features of the provided subjectmatter. Specific examples of components, values, operations, materials,arrangements, or the like, are described below to simplify the presentdisclosure. These are, of course, merely examples and are not intendedto be limiting. Other components, values, operations, materials,arrangements, or the like, are contemplated. For example, the formationof a first feature over or on a second feature in the description thatfollows may include embodiments in which the first and second featuresare formed in direct contact, and may also include embodiments in whichadditional features may be formed between the first and second features,such that the first and second features may not be in direct contact. Inaddition, the present disclosure may repeat reference numerals and/orletters in the various examples. This repetition is for the purpose ofsimplicity and clarity and does not in itself dictate a relationshipbetween the various embodiments and/or configurations discussed.

Further, spatially relative terms, such as “beneath,” “below,” “lower,”“above,” “upper” and the like, may be used herein for ease ofdescription to describe one element or feature's relationship to anotherelement(s) or feature(s) as illustrated in the figures. The spatiallyrelative terms are intended to encompass different orientations of thedevice in use or operation in addition to the orientation depicted inthe figures. The apparatus may be otherwise oriented (rotated 90 degreesor at other orientations) and the spatially relative descriptors usedherein may likewise be interpreted accordingly.

In some embodiments, a transmitter circuit includes an amplifier thatoutputs an RF signal on an output node, a power detection circuitcoupled with the output node, a reference voltage generator, acomparator that receives an output voltage of the power detectioncircuit and a reference voltage of the reference voltage generator, andan ADC coupled between the comparator and the amplifier. The powerdetection circuit generates the output voltage having a first componentbased on a DC bias voltage and a second component based on a power levelof the RF signal. The reference voltage generator generates thereference voltage based on the DC bias voltage, and the amplifieradjusts the power level of the RF signal responsive to an output of theADC.

FIG. 1 is a diagram of a transmitter circuit 100, in accordance withsome embodiments. Transmitter circuit 100 includes an amplifier 110, apower detection circuit 120, a reference voltage generator 130, acomparator 140, and an analog-to-digital converter (ADC) 150. Amplifier110 is electrically coupled to power detection circuit 120 by a node115, power detection circuit 120 is electrically coupled to comparator140 by a node 125, reference voltage generator 130 is electricallycoupled to comparator 140 by a node 135, comparator 140 is electricallycoupled to ADC 150 by a node 145, and ADC 150 is electrically coupled toamplifier 110 by a signal path 155.

In some embodiments, transmitter circuit 100 is a component of an RFtransmitter. In some embodiments, transmitter circuit 100 is a componentof an RF transmitter having an operating frequency from 2.2 gigahertz(GHz) to 2.6 GHz. In some embodiments, transmitter circuit 100 is acomponent of an RF transmitter having an operating frequency from 5.6GHz to 6.0 GHz.

In some embodiments, transmitter circuit 100 is a component of an RFtransmitter conforming to a wireless network standard. In someembodiments, transmitter circuit 100 is a component of an RF transmitterconforming to a Bluetooth standard.

Amplifier 110 is an RF power amplifier configured to receive one or moreinput signals (not shown) on one or more input terminals (not shown) andoutput a signal RFOUT on node 115. Amplifier 110 is configured togenerate signal RFOUT by increasing an amplitude of the one or moreinput signals by a gain having a gain level, and to adjust the gainlevel in response to a plurality of bits n received on signal path 155.

In some embodiments, amplifier 110 is configured to generate signalRFOUT by increasing the amplitude of an input signal relative to aground or other reference level. In some embodiments, amplifier 110 isconfigured to generate signal RFOUT by increasing the amplitude of adifferential input signal.

By adjusting the gain level in response to plurality of bits n,amplifier 110 is capable of outputting signal RFOUT having apredetermined power level controlled by plurality of bits n. In someembodiments, plurality of bits n ranges from a lowest value to a highestvalue, and amplifier 110 is configured to adjust the gain levelinversely to the value of plurality of bits n. In some embodiments,plurality of bits n ranges from a lowest value to a highest value, andamplifier 110 is configured to adjust the gain level proportionally tothe value of plurality of bits n.

In some embodiments, amplifier 110 is configured to respond to pluralityof bits n on signal path 155 having a number ranging from 4 bits to 6bits. In some embodiments, amplifier 110 is configured to respond toplurality of bits n on signal path 155 having fewer than 4 bits. In someembodiments, amplifier 110 is configured to respond to plurality of bitsn on signal path 155 having greater than 6 bits.

In some embodiments, amplifier 110 is configured to respond to pluralityof bits n received in series. In some embodiments, signal path 155includes more than one conductive path, and amplifier 110 is configuredto respond to plurality of bits n received in parallel.

Power detection circuit 120 is configured to receive signal RFOUT onnode 115 and to output a voltage VRF on node 125. Power detectioncircuit 120 is configured to generate voltage VRF having a directcurrent (DC) voltage level that includes a first component summed with asecond component. In some embodiments, voltage VRF includes one or morealternating current (AC) or other time-varying components that are smallrelative to the first and second components of the DC voltage level.

The first component of the DC voltage level corresponds to apredetermined voltage level, and the second component of the DC voltagelevel varies based on an amplitude of signal RFOUT.

In some embodiments, power detection circuit 120 is configured togenerate the second component of the DC voltage level having a magnitudethat increases with respect to an increasing amplitude, or power level,of signal RFOUT. In some embodiments, power detection circuit 120 isconfigured to generate the second component of the DC voltage levelhaving a magnitude that decreases with respect to an increasingamplitude, or power level, of signal RFOUT.

Reference voltage generator 130 is configured to generate a referencevoltage VREF and output reference voltage VREF on node 135. Referencevoltage generator 130 is configured to generate reference voltage VREFhaving a DC voltage level corresponding to the first component of the DCvoltage level of voltage VRF.

In some embodiments, reference voltage generator 130 is configured togenerate reference voltage VREF having a DC voltage level equal to thefirst component of the DC voltage level of voltage VRF. In someembodiments, reference voltage generator 130 is configured to generatereference voltage VREF having a DC voltage level equal to the firstcomponent of the DC voltage level of voltage VRF plus or minus an offsetvalue. In some embodiments, reference voltage generator 130 isconfigured to generate reference voltage VREF having a DC voltage levelproportional to the first component of the DC voltage level of voltageVRF.

Comparator 140 is configured to receive voltage VRF on node 125, receivereference voltage VREF on node 135, generate a voltage VCMP based on adifference between voltage VRF and reference voltage VREF, and outputvoltage VCMP on node 145. In some embodiments, comparator 140 isconfigured to generate voltage VCMP equal to a difference betweenvoltage VRF and reference voltage VREF.

Because reference voltage VREF corresponds to the first component of theDC level of voltage VRF, and voltage VCMP is based on the differencebetween voltage VRF and reference voltage VREF, voltage VCMP has amagnitude that varies based on the second component of the DC level ofvoltage VRF, and therefore based on the amplitude, or power level, ofsignal RFOUT.

ADC 150 is configured to receive voltage VCMP on node 145 and generateplurality of bits n on signal path 155 based on voltage VCMP. In someembodiments, ADC 150 is configured to generate plurality of bits nranging from a lowest value to a highest value, with the value varyingproportionally to a value of voltage VCMP. In some embodiments, ADC 150is configured to generate plurality of bits n ranging from a lowestvalue to a highest value, with the value varying inversely to a value ofvoltage VCMP. In some embodiments, ADC 150 is a successive approximationregister (SAR) ADC.

In some embodiments, ADC 150 is configured to generate plurality of bitsnon signal path 155 having a number ranging from 4 bits to 6 bits. Insome embodiments, ADC 150 is configured to generate plurality of bits non signal path 155 having fewer than 4 bits. In some embodiments, ADC150 is configured to generate plurality of bits n on signal path 155having greater than 6 bits.

In some embodiments, ADC 150 is configured to generate plurality of bitsnon signal path 155 in series. In some embodiments in which signal path155 includes more than one conductive path, ADC 150 is configured togenerate plurality of bits n on signal path 155 in parallel.

In operation, plurality of bits n, based on voltage VCMP, providefeedback to amplifier 110 such that adjustments to the gain level ofamplifier 110 maintain a predetermined power level of signal RFOUT. Insome embodiments, in operation, amplifier 110 increases the gain levelin response to plurality of bits n having one or more first values anddecreases the gain level in response to plurality of bits n having oneor more second values different from the one or more first values. Insome embodiments, in operation, amplifier 110 adjusts the gain level inresponse to plurality of bits n having one or more first values andmaintains a previous gain level in response to plurality of bits nhaving one or more second values different from the one or more firstvalues.

In some embodiments, in operation, amplifier 110 sets the gain level toa level of a plurality of levels, each level corresponding to a value ofa plurality of values of plurality of bits n.

By this configuration, transmitter circuit 100 is capable ofcompensating for process and temperature variations in an output powerof amplifier 110. The comparator-based architecture of transmittercircuit 100 enables process and temperature variation compensation withlower power and smaller area requirements compared to other approachesfor making amplifier gain adjustments.

In some embodiments, amplifier 110, power detection circuit 120,reference voltage generator 130, comparator 140, and ADC 150 arecomponents of a single IC chip, and transmitter circuit 100 uses feweroff-chip components and less space than are required using otherapproaches for making amplifier gain adjustments.

FIG. 2 is a diagram of a power detection circuit 200, in accordance withsome embodiments. Power detection circuit 200 is usable as powerdetection circuit 120, described above with respect to transmittercircuit 100 and FIG. 1. Power detection circuit 200 includes devices C1,R1, M1, M2, M3, and C2. Power detection circuit 200 also includes aninput node IN configured to receive an RF signal RFIN, and an outputnode OUT. In some embodiments, RF signal RFIN is RF signal RFOUT, inputnode IN is node 115, and output node OUT is node 125, each describedabove with respect to transmitter circuit 100 and FIG. 1.

Device C1 is coupled between input node IN and an internal node INT1.Device R1 is coupled between an input node BIAS and internal node INT1.Device M1 is coupled between internal node INT1, a signal node CLK, anda ground node GND. Device M2 is coupled between internal node INT1, apower node PWR, and output node OUT. Device M3 is coupled between outputnode OUT, signal node CLK, and ground node GND. Device C2 is coupledbetween output node OUT and ground node GND.

Power detection circuit 200 is configured to switch between a first modeof operation and a second mode of operation responsive to a signal ϕCLKon signal node CLK, as described below.

Device C1 is a circuit component configured to receive RF signal RFIN oninput node IN and transfer at least a portion of RF signal RFIN tointernal node INT1 by capacitively coupling input node IN to internalnode INT1.

In the embodiment depicted in FIG. 2, device C1 is a capacitive element.In some embodiments, device C1 is a capacitor. In some embodiments,device C1 is a transistor or other suitable device configured totransfer at least a portion of RF signal RFIN from input node IN tointernal node INT1.

Device R1 is a circuit component configured to transfer at least aportion of a DC voltage VBIAS from input BIAS to internal node INT1 byresistively coupling input node BIAS to internal node INT1. In theembodiment depicted in FIG. 2, device R1 is a resistive element. In someembodiments, device R1 is a resistor. In some embodiments, device R1 isa transistor or other suitable device configured to transfer at least aportion of DC voltage VBIAS from input node BIAS to internal node INT1.

Device M1 is a circuit component configured to selectively coupleinternal node INT1 to ground node GND, responsive to signal ϕCLK onsignal node CLK.

In the first mode of operation, device M1 responds to a first logicalstate of signal ϕCLK by coupling internal node INT1 to ground node GNDwith a low-impedance path for both AC and DC components of voltage VIN.In the second mode of operation, device M1 responds to a second logicalstate of signal ϕCLK different from the first state by capacitivelycoupling internal node INT1 to ground node GND, thereby providing ahigh-resistance path for a DC component of voltage VIN.

In the embodiment depicted in FIG. 2, device M1 is an n-type metal oxidesemiconductor (NMOS) transistor configured to operate in the first modein response to signal ϕCLK having a high voltage value corresponding tothe first logical state and to operate in the second mode in response tosignal ϕCLK having a low voltage value corresponding to the secondlogical state. In some embodiments, device M1 is a p-type metal oxidesemiconductor (PMOS) transistor or other suitable device configured toselectively operate in first and second modes in response to signalϕCLK, as described above.

By the configuration of devices R1 and M1, DC voltage VBIAS is dividedto generate a DC node voltage component of voltage VIN on internal nodeINT1. In the first mode of operation, because device M1 provides alow-resistance path between internal node INT1 and ground node GND, theDC node voltage component of voltage VIN has a value at or near a groundvoltage level on ground node GND.

In the second mode of operation, because device M1 provides ahigh-resistance path between internal node INT1 and ground node GND, theDC node voltage has a value between a value of DC voltage VBIAS and theground voltage level. The value of the DC node voltage component ofvoltage VIN is determined by the relative resistance values of devicesR1 and M1 while operating in the second mode of operation.

By the configuration of devices C1 and M1, RF signal RFIN is divided togenerate an RF component of voltage VIN on internal node INT1. In thefirst mode of operation, because device M1 provides a low-impedance pathbetween internal node INT1 and ground node GND, the RF component ofvoltage VIN has a value at or near the ground voltage level on groundnode GND.

In the second mode of operation, because device M1 capacitively couplesinternal node INT1 to ground node GND, devices C1 and M1 divide RFsignal RFIN to generate an RF modulation signal as the AC component ofvoltage VIN. The RF modulation signal has a magnitude that is a fractionof the magnitude of RF signal RFIN. The ratio of the magnitude of the RFmodulation signal to the magnitude of RF signal RFIN is determined bythe relative capacitance values of devices C1 and M1 while operating inthe second operating mode.

Devices C1, R1, and M1 are thereby configured to generate voltage VINhaving DC and AC components at or near the ground voltage level in thefirst mode of operation, and to generate voltage VIN in the second modeof operation having a DC node voltage that is a fraction of DC voltageVBIAS and an RF modulation signal that is a fraction of RF signal RFIN.

The fractional value of the DC node voltage results from the voltagedivider configuration of a resistance value RR1 of device R1 and aresistance value RM1 of the high-resistance path of device M1 whileoperating in the second mode of operation, and is therefore determinedby the equation:DC node voltage value=VBIAS×RM1/(RM1+RR1).  (1)

The fractional value of the RF modulation signal results from thevoltage divider configuration of a capacitance value CC1 of device C1and a capacitance value CM1 of device M1 while operating in the secondmode of operation, and is therefore determined by the equation:RF modulation signal value=RFIN×CC1/(CM1+CC1).  (2)

Device M2 is a circuit component configured to selectively transfer, ortrace, voltage VIN from internal node INT1 to output node OUT. Device M2is configured to receive a power supply voltage VDD on power node PWRand to selectively output current ID on output node OUT.

In the embodiment depicted in FIG. 2, device M2 is an NMOS transistorconfigured as a source follower. In some embodiments, device M2 is aPMOS transistor or other suitable device configured to selectivelytransfer voltage VIN from internal node INT1 to output node OUT and toselectively output current ID on output node OUT.

Device M3 is a circuit component configured to selectively couple outputnode OUT to ground node GND, responsive to signal ϕCLK on signal nodeCLK.

In the first mode of operation, device M3 responds to the first logicalstate of signal ϕCLK by coupling output node OUT to ground node GND witha low-impedance AC and DC path. In the second mode of operation, deviceM3 responds to the second logical state of signal ϕCLK by capacitivelycoupling output node OUT to ground node GND, thereby providing ahigh-resistance path for the DC node voltage component of voltage VRF onoutput node OUT. Device M3 is configured so that, in the second mode ofoperation, the capacitive coupling between output node OUT and groundnode GND is asymmetrical such that the RF modulation signal component ofvoltage VRF is rectified.

In the embodiment depicted in FIG. 2, device M3 is an NMOS transistorconfigured to operate in the first mode in response to signal ϕCLKhaving a high voltage value corresponding to the first logical state andto operate in the second mode in response to signal ϕCLK having a lowvoltage value corresponding to the second logical state. In someembodiments, device M3 is a PMOS transistor or other suitable deviceconfigured to selectively operate in first and second modes in responseto signal ϕCLK, as described above.

Device C2 is a circuit component configured to capacitively coupleoutput node OUT to ground node GND. In the embodiment depicted in FIG.2, device C2 is a capacitive element. In some embodiments, device C2 isa capacitor. In some embodiments, device C2 is a transistor or othersuitable device configured to capacitively couple output node OUT toground node GND.

In the first mode of operation, voltage VIN having DC and AC componentsat or near the ground voltage level, as described above, causes deviceM2 to be switched off and current ID to be at or near zero. Becausedevice M3 is also configured to couple output node OUT to ground nodeGND with a low-impedance AC and DC path in the first mode of operation,voltage VRF is generated on output node OUT having a value at or nearthe ground voltage level.

In the second mode of operation, because device M2 is configured as asource follower and both of devices M3 and C2 are configured ashigh-resistance DC paths, the DC node voltage component of voltage VINis transferred to output node OUT to generate a first DC component ofvoltage VRF.

In the second mode of operation, devices M3 and C2 act to rectify the RFmodulation signal component of voltage VIN that is transferred frominternal node INT1 to output node OUT. The RF modulation signalcomponent of voltage VIN on internal node INT1 is therefore transferredto output node OUT as a second DC component of voltage VRF.

In the second mode of operation, because the RF modulation signalcomponent of voltage VIN is generated by dividing RF signal RFIN, asdescribed above, the magnitude of the second DC component of voltage VRFon output node OUT increases with respect to an increasing magnitude ofRF signal RFIN, and decreases with respect to a decreasing magnitude ofRF signal RFIN.

Power detection circuit 200 is configured to generate the DC componentof voltage VIN by dividing DC voltage VBIAS; a sensitivity of a transferfunction of power detection circuit 200 is therefore capable of beingadjusted by adjusting the voltage level of DC voltage VBIAS. Becausedevice M3 is switched on in the second mode of operation, powerdetection circuit 200 is configured to respond to DC voltage VBIAShaving a voltage level that ranges from a threshold voltage of device M3to power supply voltage VDD.

As described above, power detection circuit 200 is configured togenerate voltage VRF having a first DC component based on DC voltageVBIAS and a second DC component based on the magnitude, or power level,of RF signal RFIN. Power detection circuit 200 is thereby capable ofgenerating the first and second DC components of voltage VRF fordetecting a power level of RF signal RFIN using less power than otherapproaches for detecting a power level of an RF signal.

FIG. 3 is a diagram of a reference voltage generator 300, in accordancewith some embodiments. Reference voltage generator 300 is usable asreference voltage generator 130, described above with respect totransmitter circuit 100 and FIG. 1. Reference voltage generator 300includes devices R2, M4, M5, M6, and C3. Reference voltage generator 300also includes an output node REF. In some embodiments, output node REFis node 135 described above with respect to transmitter circuit 100 andFIG. 1.

Device R2 is coupled between input node BIAS and an internal node INT2.Device M4 is coupled between internal node INT2, signal node CLK, andground node GND. Device M5 is coupled between internal node INT2, powernode PWR, and output node REF. Device M6 is coupled between output nodeREF, signal node CLK, and ground node GND. Device C3 is coupled betweenoutput node REF and ground node GND.

Devices R2, M4, M5, M6, and C3 correspond to devices R1, M1, M2, M3, andC2, respectively, of power detection circuit 200, described above withrespect to FIG. 2, and reference voltage generator 300 has aconfiguration that matches the configuration of power detection circuit200, with the exception of a device that corresponds to device C1 ofpower detection circuit 200.

Accordingly, reference voltage generator 300 is configured to generateoutput voltage VREF on output node REF having a voltage level thatincludes the first DC component of voltage VRF but does not include thesecond DC component of voltage VRF, described above with respect topower detection circuit 200 and FIG. 2.

Similar to power detection circuit 200, reference voltage generator 300is configured to switch between a first mode of operation and a secondmode of operation responsive to signal ϕCLK on signal node CLK. Duringthe first mode of operation, voltage generator 300 generates voltageVREF on output node REF having a value at or near the ground voltagelevel. During the second mode of operation, voltage generator 300generates voltage VREF on output node REF having the value of the firstDC component of voltage VRF.

Because both power detection circuit 200 and reference voltage generator300 are configured to switch between first and second modes of operationresponsive to the same signal ϕCLK, power detection circuit 200 andreference voltage generator 300, in operation, switch between first andsecond modes of operation synchronously.

In some embodiments, at least one of power detection circuit 200 orreference voltage generator 300 is configured to respond to signal ϕCLKbeing a clock signal. In some embodiments, at least one of powerdetection circuit 200 or reference voltage generator 300 is configuredto respond to signal ϕCLK having a switching frequency of 100 Hertz (Hz)to 1000 Hz. In some embodiments, at least one of power detection circuit200 or reference voltage generator 300 is configured to respond tosignal ϕCLK having a switching frequency below 100 Hz. In someembodiments, at least one of power detection circuit 200 or referencevoltage generator 300 is configured to respond to signal ϕCLK having aswitching frequency above 1000 Hz.

FIG. 4 is a diagram of a comparator 400, in accordance with someembodiments. Comparator 400 is usable as comparator 140, described abovewith respect to transmitter circuit 100 and FIG. 1. Comparator 400includes transistors M1A, M2A, M1B, and M2B coupled between power nodePWR, carrying power supply voltage VDD, and a node INT3, and transistorMBIAS coupled between node INT3 and ground node GND, at the groundvoltage level.

Comparator 400 also includes an input node INP, an input node INN, andan output node CMPOUT. In some embodiments, input node INP is node 125,input node INN is node 135, and output node CMPOUT is node 145, eachdescribed above with respect to transmitter circuit 100 and FIG. 1.

Transistors M2A and M2B are PMOS transistors having source terminalscoupled to power node PWR and gate terminals coupled to each other andto a drain terminal of transistor M2A. Transistors M2A and M2B arethereby configured as a current mirror such that, in operation, acurrent IM2A flowing through transistor M2A has a same value as that ofa current IM2B flowing through transistor M2B.

Transistor MIA is an NMOS transistor having a source terminal coupled tonode INT3, a drain terminal coupled to the drain terminal of transistorM2A, and a gate terminal coupled to node INP carrying voltage VRF.Transistor MIA is thereby configured in series with transistor M2A suchthat, in operation, current IM2A flows through transistor MIA.

Transistor M1B is an NMOS transistor having a source terminal coupled tonode INT3, a drain terminal coupled to the drain terminal of transistorM2B and node CMPOUT, and a gate terminal coupled to node INN carryingvoltage VREF. Node CMPOUT is configured so that, in operation, currenton node CMPOUT is not significant. Transistor M1B is thereby configuredin series with transistor M2B such that, in operation, substantially allof current IM2B flows through transistor M1B.

Because current IM2A is mirrored to current IM2B, in operation, currentsthrough transistors MIA and M1B are the same. In operation, voltage VRFat the gate terminal of transistor M1A, by controlling current flowthrough transistor M1A, also controls current flow through transistorM1B.

In operation, transistor M1B is controlled both by voltage VREF at thegate terminal of transistor M1B and by voltage VRF at the gate terminalof transistor M1A. By the configuration of transistors MIA and M1B, inoperation, a voltage drop across the drain and source terminals oftransistor M1B increases as voltage VRF increases relative to voltageVREF and decreases as voltage VRF decreases relative to voltage VREF.Thus, transistors MIA and M1B are referred to as a differential pair.

Transistor MBIAS is an NMOS transistor having a drain terminal coupledto node INT3, a source terminal coupled to ground node GND, and a gateterminal coupled to node BIAS carrying DC voltage VBIAS. In operation, avoltage on node INT3 relative to the ground voltage level is controlledby a value of DC voltage VBIAS at the gate terminal of transistor MBIAS.

Voltage VCMP on node CMPOUT is the sum of the voltage drop acrosstransistor M1B and the voltage on node INT3. For a given value of DCvoltage VBIAS, voltage VCMP is controlled by the differential values ofvoltages VRF and VREF. Comparator 400 is thereby configured as anamplifier such that, in operation, comparator 400 receives voltage VRFon node INP, receives voltage VREF on node INN, generates voltage VCMPbased on voltage VREF subtracted from voltage VRF, and outputs voltageVCMP on output node CMPOUT.

Comparator 400 is configured to receive DC voltage VBIAS on node BIASand to adjust a sensitivity of the differential pair of transistors MIAand M1B to voltages VRF and VREF. In operation, a sufficiently largevalue of DC voltage VBIAS, e.g., VDD, causes transistor MBIAS to befully on such that the voltage on node INT3 is near the ground voltagelevel. As DC voltage VBIAS decreases in value, the voltage on node INT3and the source terminals of transistors MIA and M1B increases. Becausetransistors MIA and M1B are controlled by voltages VRF and VREF asdiscussed above, increasing the voltage on node INT3 increases thesensitivity of transistors MIA and M1B to changes in the values ofvoltages VRF and VREF at the gate terminals of transistors MIA and M1B.

In addition to comparator 400, both power detection circuit 200 andreference voltage generator 300 are configured to receive DC voltageVBIAS, as discussed above. In some embodiments in which transmittercircuit 100 includes power detection circuit 200, reference voltagegenerator 300, and comparator 440, transmitter circuit 100 is therebyconfigured to, in operation, adjust an overall sensitivity to the powerlevel of signal RFOUT.

FIG. 5 is a timing diagram of voltage and current signals at variousnodes of power detection circuit 200, in accordance with someembodiments. FIG. 5 depicts voltages VIN and VRF, current ID, and signalϕCLK, described above with respect to detection circuit 200 and FIG. 2.

At a time t1, signal ϕCLK switches from a high logical level to a lowlogical level, and, at a time t2, signal ϕCLK switches from the lowlogical level to the high logical level.

At time t1, voltage VIN rises asymptotically from a value at or near theground voltage level to a second value that includes the RF modulationsignal superimposed on the DC node voltage component of voltage VIN, asdescribed above. At time t2, voltage VIN returns to the value at or nearthe ground voltage level.

At time t1, voltage VRF rises asymptotically from a value at or near theground voltage level to a second value that includes the first DCcomponent traced from the DC node voltage component of voltage VIN andthe second DC component generated from the rectified RF modulationsignal on node INT1, as described above. At time t2, voltage VRF returnsto the value at or near the ground voltage level.

At time t1, current ID increases transiently as output node OUT ischarged to the second value of voltage VRF, then decays to a value thatessentially includes an RF component having an average at or near theground voltage level.

FIG. 6 is a diagram of a transfer function of power detection circuit200, in accordance with some embodiments. FIG. 6 depicts voltage VRF asa function of the amplitude of RF signal RFIN, described above withrespect to detection circuit 200 and FIG. 2. In the embodiment depictedin FIG. 6, voltage VRF increases linearly with respect to an increasingamplitude of RF signal RFIN.

FIGS. 7A and 7B are timing diagrams of voltage signals at various nodesof transmitter circuit 100, in accordance with some embodiments. Each ofFIGS. 7A and 7B depicts voltages VIN, VRF, VREF, and VCMP, and signalϕCLK, described above with respect to transmitter circuit 100 andFIG. 1. FIG. 7A depicts a first case in which signal RFOUT has a firstamplitude, and FIG. 7B depicts a second case in which signal RFOUT has asecond amplitude larger than the first amplitude.

In both cases, between time t1, at which point signal ϕCLK switches fromthe high logical level to the low logical level, and time t2, at whichpoint signal ϕCLK switches from the low logical level to the highlogical level, each of signals VIN, VRF, and VREF rises asymptoticallyto a final value.

In response to the relative magnitudes of signal RFOUT, the RF componentof voltage VIN depicted in FIG. 7A is smaller than the RF component ofvoltage VIN depicted in FIG. 7B. The DC value of voltage VRF isaccordingly smaller in the first case depicted in FIG. 7A as compared tothe second case depicted in FIG. 7B. Because voltage VREF is independentof the magnitude of signal RFOUT, voltage VREF is unchanged in thedepictions of FIGS. 7A and 7B.

Because of the increase in value of voltage VRF in response to theincreased amplitude of signal RFOUT, the value of voltage VCMP in thesecond case, depicted in FIG. 7B, is larger than the value of VCMP inthe first case, depicted in FIG. 7A.

In operation, plurality of bits n generated by ADC 150 from voltage VCMPtherefore represent the relative magnitude of signal RFOUT, and are usedto make gain adjustments to amplifier 110 accordingly.

As described above with respect to power detection circuit 200 and FIG.5, current ID has an average value at or near the ground reference valuewhile outputting voltage VRF indicative of the power of signal RFOUT. Asdescribed above with respect to comparator 400 and FIGS. 7A and 7B,voltages VRF and VREF are received at high-impedance gate terminals ofcomparator 400 while outputting voltage VCMP indicative of the power ofsignal RFOUT.

By including power detection circuit 200, transmitter circuit 100 iscapable of generating an indication of the amplitude of signal RFOUTwhile using current ID having a small value, thereby operating at a lowpower level. By including similarly configured reference voltagegenerator 300, transmitter circuit 100 is capable of generating areference voltage while operating at a low power level. Transmittercircuit 100 is thereby configured to calibrate the power of outputsignal RFOUT using less power and DC current than are required in otherRF signal calibration approaches. In some embodiments, transmittercircuit 100 is configured to calibrate signal RFOUT using a DC currentthat ranges from 50 microamperes (μA) to 100 μA.

In some embodiments, because power detection circuit 200 and referencevoltage generator 300 are configured to operate in a first mode ofoperation in which voltages VRF and VREF are generated having values ator near the ground voltage level, power consumption averaged over thefirst and second modes of operation is further reduced in comparison toother approaches.

FIG. 8 is a flowchart of a method 800 of method of detecting a powerlevel of an RF signal, in accordance with one or more embodiments. Insome embodiments, method 800 is implemented to detect a power level ofsignal RFOUT of transmitter circuit 100, described above with respect toFIG. 1.

In some embodiments, operations in addition to those depicted in FIG. 8are performed before, between, and/or after the operations depicted inFIG. 8. In some embodiments, the operations depicted in FIG. 8 areperformed in an order other than the order depicted in FIG. 8.

At operation 810, in some embodiments, an RF signal is generated usingan amplifier. The RF signal has an amplitude that corresponds to a powerlevel of the RF signal. In some embodiments, generating the RF signalincludes generating an RF signal using a power amplifier of an RFtransmitter circuit. In some embodiments, generating the RF signalincludes generating signal RFOUT using amplifier 110, described abovewith respect to transmitter circuit 100 and FIG. 1.

At operation 820, an internal node of a power detection circuit isdriven to a DC node voltage level. In some embodiments, driving theinternal node of a power detection circuit to a DC node voltage levelincludes dividing an input voltage with a voltage divider. In someembodiments, driving the internal node of a power detection circuit to aDC node voltage level includes driving the internal node of the powerdetection circuit responsive to a clock signal. In some embodiments,driving the internal node of a power detection circuit to a DC nodevoltage level includes driving internal node INT1 to the DC node voltagecomponent of voltage VIN, described above with respect to powerdetection circuit 200 and FIG. 2.

At operation 830, from the DC node voltage level, a first component ofan output voltage is generated on an output node of the power detectioncircuit. In some embodiments, generating the first component of theoutput voltage includes generating a first DC component of the outputvoltage. In some embodiments, generating the first component of theoutput voltage includes tracing the DC node voltage level using a sourcefollower.

In some embodiments, generating the first component of the outputvoltage includes generating voltage VRF on node 125 using powerdetection circuit 120, described above with respect to transmittercircuit 100 and FIG. 1. In some embodiments, generating the firstcomponent of the output voltage includes generating the first DCcomponent of voltage VRF on output node OUT, described above withrespect to power detection circuit 200 and FIG. 2.

At operation 840, the RF signal is received on an input node of thepower detection circuit. In some embodiments, receiving the RF signalincludes receiving the RF signal with a capacitive element coupled tothe input node of the power detection circuit. In some embodiments,receiving the RF signal includes receiving signal RFIN on input node IN,described above with respect to power detection circuit 200 and FIG. 2.

At operation 850, the RF signal is divided to generate a modulationsignal on the internal node of the power detection circuit. In someembodiments, dividing the RF signal to generate the modulation signalincludes dividing the RF signal responsive to a clock signal. In someembodiments, dividing the RF signal to generate the modulation signalincludes dividing the RF signal using one or more devices used togenerate the DC node voltage level. In some embodiments, dividing the RFsignal to generate the modulation signal includes dividing RF signalRFIN to generate the RF modulation signal component of voltage VIN oninternal node INT1, described above with respect to power detectioncircuit 200 and FIG. 2.

At operation 860, by at least partially rectifying the modulationsignal, a second component of the output voltage is generated on theoutput node. In some embodiments, generating the second component of theoutput voltage includes generating a second DC component of the outputvoltage. In some embodiments, generating the second component of theoutput voltage includes generating the second component of the outputvoltage in response to a clock signal.

In some embodiments, generating the second component of the outputvoltage includes generating voltage VRF on node 125 using powerdetection circuit 120, described above with respect to transmittercircuit 100 and FIG. 1. In some embodiments, generating the secondcomponent of the output voltage includes generating the second DCcomponent of output voltage VRF on output node OUT, described above withrespect to power detection circuit 200 and FIG. 2.

At operation 870, in some embodiments, a reference voltage is generatedon a reference circuit output node. In some embodiments, generating thereference voltage includes generating the reference voltage having asame value as a value of the first component of the output voltage.

In some embodiments, generating the reference voltage includesgenerating voltage VREF on node 135 using reference voltage generator130, described above with respect to transmitter circuit 100 and FIG. 1.In some embodiments, generating the reference voltage includesgenerating voltage VREF on output node REF, described above with respectto reference voltage generator 300 and FIG. 3.

At operation 880, in some embodiments, a comparison voltage is generatedbased on a difference between the output voltage and the referencevoltage. In some embodiments, generating the comparison voltage includesgenerating the comparison voltage equal to the difference between theoutput voltage and the reference voltage.

In some embodiments, generating the comparison voltage includesgenerating voltage VCMP on node 145 using comparator 140, describedabove with respect to transmitter circuit 100 and FIG. 1. In someembodiments, generating the comparison voltage includes generatingvoltage VCMP on output node CMPOUT, described above with respect tocomparator 400 and FIG. 4.

At operation 890, in some embodiments, a gain of the amplifier isadjusted based on the comparison voltage. In some embodiments, adjustingthe gain of the amplifier includes adjusting the amplifier to have again sufficient to output an RF signal having a predetermined powerlevel. In some embodiments, adjusting the gain of the amplifier includesadjusting the gain of amplifier 110 to output signal RFOUT having apredetermined power level, described above with respect to transmittercircuit 100 and FIG. 1.

In some embodiments, adjusting the gain of the amplifier includesconverting the comparison voltage to a digital signal. In someembodiments, adjusting the gain of the amplifier includes convertingvoltage VCMP to bits n on signal path 155 using ADC 150, described abovewith respect to transmitter circuit 100 and FIG. 1.

In some embodiments, adjusting the gain of the amplifier includesconverting the comparison voltage to a digital signal by initiating asuccessive approximation sequence in an ADC. In some embodiments,adjusting the gain of the amplifier includes converting the comparisonvoltage to a digital signal during a second mode of operation, describedabove with respect to power detection circuit 200, reference voltagegenerator 300, and FIGS. 2 and 3.

By executing some or all of the operations of method 800, a power levelof an RF signal is detected while obtaining the power and circuit sizebenefits described above with respect to transmitter circuit 100, powerdetection circuit 200, reference voltage generator 300, and comparator400.

In some embodiments, a method of detecting a power level of an RF signalincludes driving an internal node of a power detection circuit to a DCnode voltage level, generating, based on the DC node voltage level, afirst component of an output voltage on an output node of the powerdetection circuit, receiving the RF signal on an input node of the powerdetection circuit, dividing the RF signal to generate a modulationsignal on the internal node, and generating, by at least partiallyrectifying the modulation signal, a second component of the outputvoltage on the power detection circuit output node.

In some embodiments, a method of detecting a power level of an RF signalincludes driving an internal node of a power detection circuit to a DCnode voltage level, generating, based on the DC node voltage level, afirst component of an output voltage on an output node of the powerdetection circuit, receiving the RF signal on an input node of the powerdetection circuit, dividing the RF signal to generate a modulationsignal on the internal node, generating, by at least partiallyrectifying the modulation signal, a second component of the outputvoltage on the power detection circuit output node, generating areference voltage on a reference circuit output node, the referencevoltage having a value equivalent to the first component of the outputvoltage, and generating a comparison voltage based on a differencebetween the output voltage and the reference voltage.

In some embodiments, a method of detecting a power level of an RF signalincludes generating the RF signal using an amplifier having a gain levelresponsive to a plurality of bits, driving an internal node of a powerdetection circuit to a DC node voltage level, generating, based on theDC node voltage level, a first component of an output voltage on anoutput node of the power detection circuit, receiving the RF signal onan input node of the power detection circuit, dividing the RF signal togenerate a modulation signal on the internal node, generating, by atleast partially rectifying the modulation signal, a second component ofthe output voltage on the power detection circuit output node,generating a comparison voltage based on a difference between the outputvoltage and a reference voltage, and generating the plurality of bitsbased on the comparison voltage.

The foregoing outlines features of several embodiments so that thoseskilled in the art may better understand the aspects of the presentdisclosure. Those skilled in the art should appreciate that they mayreadily use the present disclosure as a basis for designing or modifyingother processes and structures for carrying out the same purposes and/orachieving the same advantages of the embodiments introduced herein.Those skilled in the art should also realize that such equivalentconstructions do not depart from the spirit and scope of the presentdisclosure, and that they may make various changes, substitutions, andalterations herein without departing from the spirit and scope of thepresent disclosure.

What is claimed is:
 1. A method of detecting a power level of a radiofrequency (RF) signal, the method comprising: driving an internal nodeof a power detection circuit to a direct current (DC) node voltagelevel; generating, based on the DC node voltage level, a first componentof an output voltage on an output node of the power detection circuit;receiving the RF signal on an input node of the power detection circuit;dividing the RF signal to generate a modulation signal on the internalnode; and generating, by at least partially rectifying the dividedmodulation signal, a second component of the output voltage on the powerdetection circuit output node.
 2. The method of claim 1, wherein atleast one of the driving the internal node to the DC node voltage level,the generating the first component of the output voltage, the dividingthe RF signal, or the generating the second component of the outputvoltage is performed in response to a clock signal.
 3. The method ofclaim 1, wherein the generating the second component of the outputvoltage comprises increasing a DC level of the output voltage to a levelabove the first component of the output voltage level.
 4. The method ofclaim 1, wherein the dividing the RF signal to generate the modulationsignal comprises using a capacitive element in series with a transistor.5. The method of claim 1, wherein the at least partially rectifying themodulation signal comprises using a transistor in parallel with acapacitive element.
 6. The method of claim 1, further comprising:generating the RF signal using an amplifier; generating a referencevoltage on a reference circuit output node; generating a comparisonvoltage based on a difference between the output voltage and thereference voltage; and adjusting a gain of the amplifier based on thecomparison voltage.
 7. The method of claim 1, wherein the driving theinternal node of the power detection circuit to the DC node voltagelevel comprises dividing a bias voltage using a resistive device inseries with a transistor.
 8. The method of claim 1, wherein each of thegenerating the first component of the output voltage and the generatingthe second component of the output voltage is included in a first modeof operation of the power detection circuit, and the method furthercomprises coupling the output node of the power detection circuit to aground node in a second mode of operation of the power detectioncircuit.
 9. A method of detecting a power level of a radio frequency(RF) signal, the method comprising: driving an internal node of a powerdetection circuit to a direct current (DC) node voltage level;generating, based on the DC node voltage level, a first component of anoutput voltage on an output node of the power detection circuit;receiving the RF signal on an input node of the power detection circuit;dividing the RF signal to generate a modulation signal on the internalnode; generating, by at least partially rectifying the dividedmodulation signal, a second component of the output voltage on the powerdetection circuit output node; generating a reference voltage on areference circuit output node, the reference voltage having a valueequivalent to the first component of the output voltage; and generatinga comparison voltage based on a difference between the output voltageand the reference voltage.
 10. The method of claim 9, furthercomprising: receiving a clock signal at the power detection circuit andat a reference circuit comprising the reference circuit output node,wherein each of the driving the internal node to the DC node voltagelevel, the generating the first component of the output voltage, thedividing the RF signal, the generating the second component of theoutput voltage, and the generating the reference voltage is performed inresponse to a first logical value of the clock signal.
 11. The method ofclaim 10, further comprising, in response to a second logical value ofthe clock signal different from the first logical value: coupling theinternal node and the output node of the power detection circuit to aground node; and coupling the reference circuit output node to theground node.
 12. The method of claim 9, wherein the driving the internalnode of the power detection circuit to the DC node voltage levelcomprises dividing a bias voltage using a first resistive device inseries with a first transistor, and the generating the reference voltagecomprises dividing the bias voltage using a second resistive device inseries with a second transistor.
 13. The method of claim 12, wherein thegenerating the comparison voltage comprises generating the comparisonvoltage responsive to the bias voltage.
 14. The method of claim 9,wherein the generating the comparison voltage comprises: receiving theoutput voltage at a first transistor of a differential transistor pairof a comparator; receiving the reference voltage at a second transistorof the differential transistor pair of the comparator; and subtractingthe reference voltage from the output voltage.
 15. A method of detectinga power level of a radio frequency (RF) signal, the method comprising:generating the RF signal using an amplifier having a gain levelresponsive to a plurality of bits; driving an internal node of a powerdetection circuit to a direct current (DC) node voltage level;generating, based on the DC node voltage level, a first component of anoutput voltage on an output node of the power detection circuit;receiving the RF signal on an input node of the power detection circuit;dividing the RF signal to generate a modulation signal on the internalnode; generating, by at least partially rectifying the dividedmodulation signal, a second component of the output voltage on the powerdetection circuit output node; generating a comparison voltage based ona difference between the output voltage and a reference voltage; andgenerating the plurality of bits based on the comparison voltage. 16.The method of claim 15, wherein the generating the RF signal comprisesadjusting the gain level responsive to the plurality of bits.
 17. Themethod of claim 15, wherein each of the driving the internal node of thepower detection circuit and the generating the comparison voltagecomprises receiving a bias voltage.
 18. The method of claim 15, furthercomprising: receiving the comparison voltage at an analog-to-digitalconverter (ADC); and using the ADC to generate the plurality of bitsbased on the comparison voltage.
 19. The method of claim 18, wherein theusing the ADC to generate the plurality of bits comprises using asuccessive approximation register (SAR) ADC.
 20. The method of claim 15,further comprising: generating the reference voltage using a referencecircuit; and synchronously switching the power detection and referencecircuits between first and second modes of operation, wherein each ofthe generating the first component of the output voltage, the generatingthe second component of the output voltage, and the generating thereference voltage is included in the first mode of operation.